Switching power supply

ABSTRACT

A switching power supply includes a transformer having a primary winding connected to a pair of a.c. input terminals via a rectifier circuit, and a secondary winding connected to a pair of d.c. output terminals via a rectifying and smoothing circuit. Connected between the pair of outputs of the rectifier circuit via an inductor, reverse-blocking diode, and part or whole of the transformer primary, a primary switch is turned on and off at a repetition frequency higher than the frequency of the a.c. input voltage. A soft-switching circuit is provided which comprises a serial connection of a transformer tertiary, an ancillary diode and an ancillary switch. This serial circuit is connected in parallel with the serial connection of the transformer primary and primary switch for zero-voltage switching of the primary switch.

BACKGROUND OF THE INVENTION

This invention relates to electric power supplies, and particularly to a switching power supply capable of a.c. to d.c. voltage conversion, featuring provisions for attainment of closer approximation of the input current waveform to a sinusoidal wave, and a higher power factor.

A conversion from an alternating to a direct current is possible by a rectifying and smoothing circuit comprising a rectifying circuit having a diode connected to an a.c. power supply, and a smoothing capacitor connected to the rectifying circuit. This type of rectifying and smoothing circuit possesses the disadvantage, however, of a somewhat poor power factor as a result of the fact that the smoothing capacitor is charged only at or adjacent the peaks of the a.c. voltage of sinusoidal waveform. Another drawback is that it is incapable of adjustably varying the d.c. output voltage.

Japanese Unexamined Patent Publication No. 8-154379 represents an improvement of the rectifying and smoothing circuit above. It teaches a switching power supply comprising a rectifying circuit, a smoothing capacitor, a d.c.-to-d.c. converter circuit, and an inductive reactor for a higher power factor. The reactor is electrically connected between the pair of output terminals of the rectifying circuit upon closure of a switch included in the d.c.-to-d.c. converter circuit. The desired improvement in power factor is thus attained, as the current flowing through the reactor varies in amplitude in step with the a.c. input voltage.

Despite its undisputable advantages, this prior art switching power supply has proved to be not so satisfactory as can be desired in terms of power loss.

SUMMARY OF THE INVENTION

The present invention seeks to improve the switching power supply of the noted prior art type for still higher efficiency without impairment of its inherent advantages.

Briefly, the invention may be summarized as a switching power supply capable of translating a.c. voltage into d.c. voltage. Included is a transformer connected to a pair of a.c. input terminals via a rectifier circuit, and to a pair of d.c. output terminals via a rectifying and smoothing circuit. The rectifier circuit has a first and a second output, the second being grounded in the preferred embodiments disclosed herein. A smoothing capacitor is connected between a first extremity of the primary winding of the transformer and the second output of the rectifier circuit, and an inductor between the first output of the rectifier circuit and the smoothing capacitor via at least part of the transformer primary. A primary switch is connected between a second extremity of the transformer and the second output of the rectifier circuit. The primary switch is provided with soft-switching capacitance means which can take the form of either a discrete capacitor connected in parallel therewith or parasitic capacitance of its own.

The invention particularly features a soft-switching circuit which comprises an additional winding of the transformer, and an ancillary switch connected in parallel with the smoothing capacitor at least via the additional transformer winding. The ancillary switch is designed to supply to the additional transformer winding a current of sufficient magnitude to cause the transformer primary to develop a voltage that enables the soft-switching capacitance means to discharge. A switch control circuit is connected both to the primary switch for on-off control of the primary switch at a repetition frequency higher than the frequency of the a.c. input voltage, and to the ancillary switch in order to initiate conduction through the ancillary switch earlier than the beginning of each conducting period of the primary switch and to terminate conduction through the ancillary switch not later than the end of each conducting period of the primary switch.

Such being the improved construction of the switching power supply according to the invention, a current will flow through the inductor during the conducting periods of the primary switch. Improvements in power factor and input waveform are accomplished as the inductor current has an amplitude in proportion with that of the a.c. input voltage.

The conduction of the ancillary switch, on the other hand, will result in current flow through the additional transformer winding, which is a tertiary in the preferred embodiments. Then a sufficient voltage will develop across the transformer primary to cause the soft-switching capacitance means to discharge, with a consequent drop in the voltage across the primary switch. A zero-voltage turn-on of the primary switch is thus realized for reduction of switching loss and noise.

The primary switch operates both for improvements in power factor and input waveform and for d.c.-to-d.c. conversion. The objectives of improved power factor and improved input waveform in view are thus attained with little or no addition to the size or manufacturing cost of the switching power supply.

It will also be appreciated that the winding included in the soft switching circuit is incorporated with the transformer as a tertiary in the preferred embodiments. This feature also contributes to the compactness of the device.

A further feature of the invention resides in an ancillary charging circuit connected between a third output of the rectifier circuit and the smoothing capacitor. The third output of the rectifier circuit puts out substantially the same rectifier output voltage between itself and the noted second output of the rectifier circuit as that between the first and the second output thereof. The ancillary charging circuit comprises another ancillary winding (quaternary in the preferred embodiments) of the transformer. Various specific designs will be proposed for the ancillary charging circuit.

The ancillary charging circuit is well calculated to charge the smoothing capacitor to the required degree even if the current through the primary inductor, which is for improvements in power factor and input waveform, is lessened in magnitude. The smoothing capacitor is charged both via the primary inductor and via the ancillary charging circuit. As a result, the current charging the smoothing capacitor via the primary inductor can be made less by an amount equal to the current charging the smoothing capacitor via the ancillary charging circuit than if the smoothing capacitor were charged via the primary inductor only. Power loss at the primary inductor is thus decreased, and its size can be reduced.

The ancillary charging circuit may also be utilized to make higher the voltage under which the smoothing capacitor is charged. This will serve to prevent the flow of overcurrent into the smoothing capacitor via the primary inductor at or adjacent the peaks of the a.c. input voltage. The result will be the reduction of higher harmonics of the a.c. input voltage.

An additional advantage of the ancillary charging circuit is that it makes use of an ancillary winding of the transformer. The ancillary charging circuit can therefore be most simplified in construction and reduced in size.

The transformer primary is tapped in some preferred embodiments of the invention, and the primary inductor is connected between the first output of the rectifier circuit and the smoothing capacitor via part of the transformer primary. Current will then flow through the primary inductor only when its input voltage is higher than the tap voltage, resulting in less power loss at the primary inductor.

In other embodiments, however, the transformer primary is not tapped, and the primary inductor is connected between the rectifier circuit first output and the smoothing capacitor via whole of the transformer primary. This connection is recommended where higher power factor and better input waveform are more important than efficiency, cause then the current through the primary inductor is not limited by the transformer primary.

The above and other objects, features and advantages of this invention will become more apparent, and the invention itself will best be understood, from a study of the following description and appended claims, with reference had to the attached drawings showing the preferred embodiments of the invention.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a schematic electrical diagram of a first preferred form of switching power supply according to the invention;

FIG. 2 is a schematic electrical diagram, partly in block form, showing in more detail the switch control circuit included in the FIG. 1 switching power supply;

FIG. 3, consisting of (A) through (C), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 2 switch control circuit;

FIG. 4, consisting of (A) through (H), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 1 switching power supply;

FIG. 5, consisting of (A) through (E), is a series of diagrams showing on a different time scale the voltage and current waveforms appearing in various parts of the FIG. 1 device;

FIG. 6 is a view similar to FIG. 1 but showing a second preferred form of switching power supply according to the invention;

FIG. 7 is also a view similar to FIG. 1 but showing a third preferred form of switching power supply according to the invention;

FIG. 8 is also a view similar to FIG. 1 but showing a fourth preferred form of switching power supply according to the invention;

FIG. 9 is also a view similar to FIG. 1 but showing a fifth preferred form of switching power supply according to the invention;

FIG. 10, consisting of (A) through (E), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 9 device;

FIG. 11, consisting of (A) through (H), is a series of diagrams showing the voltage and current waveforms appearing in various parts of the FIG. 9 device on a different time scale;

FIG. 12 is a view similar to FIG. 1 but showing a sixth preferred form of switching power supply according to the invention; and

FIG. 13 is also a view similar to FIG. 1 but showing a seventh preferred form of switching power supply according to the invention;

FIG. 14 is also a view similar to FIG. 1 but showing an eighth preferred form of switching power supply according to the invention;

FIG. 15 is also a view similar to FIG. 1 but showing a ninth preferred form of switching power supply according to the invention;

FIG. 16 is also a view similar to FIG. 1 but showing a tenth preferred form of switching power supply according to the invention;

FIG. 17 is also a view similar to FIG. 1 but showing an eleventh preferred form of switching power supply according to the invention;

FIG. 18 is also a view similar to FIG. 1 but showing a twelfth preferred form of switching power supply according to the invention;

FIG. 19 is also a view similar to FIG. 1 but showing a thirteenth preferred form of switching power supply according to the invention; and

FIG. 20 is also a view similar to FIG. 1 but showing a fourteenth preferred form of switching power supply according to the invention;

DESCRIPTION OF THE PREFERRED EMBODIMENTS

The switching power supply shown in FIG. 1 by way of a preferable embodiment of the invention has a pair of input terminals 1 and 2 which are to be connected to a source, not shown, of commercial a.c. voltage V_(ac) with a frequency of, for instance, 50 Hz. A noise filter 3 is connected to this pair of input terminals 1 and 2. The noise filter 3 can be of the conventional make comprising inductors and capacitors for removal of high-frequency noise from the incoming fixed-frequency alternating current.

The noise filter 3 is connected to a rectifier circuit 4 having four diodes D₁, D₂, D₃ and D₄. The first diode D₁ has its anode connected to the cathode of the second diode D₂, and its connected to the cathode of the third diode D₃. The anode of the second diode D₂ is connected to that of the fourth diode D₄. The anode of the third diode D₃ is connected to the cathode of the fourth diode D₄. The noise filter 3 has a first output conductor 41 connected to a junction 46 between the first and second diodes D₁ and D₂, and a second output conductor 42 connected to a junction 47 between the third and fourth diodes D₃ and D₄.

The rectifier circuit 4 has two output conductors 43 and 45. The first output conductors 43 is connected to a junction 48 between the cathodes of the first and third diodes D₁ and D₃. The second output conductor 45 is connected to a junction 49 between the anodes of the second and fourth diodes D₂ and D₄. It will be observed that the second rectifier output conductor 45 is grounded and will therefore be sometimes referred to as such.

At 5 is shown a transformer having a primary winding N₁, a secondary winding N₂, and, according to a feature of this invention, a tertiary or ancillary winding N₃, all wound around a magnetic core 9 and electro-magnetically coupled together. The transformer primary N₁ is tapped at 10 and thereby divided into two parts N_(1a) and N_(1b). The three transformer windings N₁, N₂ and N₃ are polarized as marked with the dots in FIG. 1. It will be seen from the markings that the transformer primary N₁ and secondary N₂ are opposite in polarity.

A smoothing capacitor C₁, preferably an electrolytic capacitor, is connected between one extremity of the transformer primary N₁ and the grounded second output conductor 45 of the rectifier circuit 4. An inductor L₁ has one extremity thereof connected to the first rectifier output conductor 43, and the other extremity to the tap 10 on the transformer primary N₁, so that the inductor is connected to the smoothing capacitor C₁ via the first part N_(1a) of the transformer primary N₁. The position of the tap 10 on the transformer primary N₁ is variable as required or desired. Generally, the smaller the ratio of the transformer primary first part N_(1a) to the transformer primary second part N_(1b), the higher will this power supply be in efficiency, but, at the same time, the less in power factor.

Shown as an insulated-gate field-effect transistor, a primary switch Q₁ is connected between the other extremity of the transformer primary N₁ and the second rectifier output conductor 45. The smoothing capacitor C₁ is in parallel with this primary switch Q₁ via the transformer primary N₁.

Also connected in parallel with the primary switch Q₁ is a soft-switching capacitor C_(q1) which is less in capacitance than the smoothing capacitor C₁. Although the soft-switching capacitor C_(q1) is shown as a discrete unit, the functions intended therefor could be served by the parasitic capacitance between the drain and source of the primary switch Q₁.

A diode D_(q1) is connected reversely in parallel with the primary switch Q₁ for its protection. The showing of this switch protection diode D_(q1) as a discrete unit is also by way of example only. The functions intended for this diode could be served by the so-called body diode of the primary switch Q₁.

The transformer secondary N₂ has its opposite extremities connected respectively to the pair of output terminals 11 and 12 via a rectifying and smoothing circuit 6. The rectifying and smoothing circuit 6 comprises a rectifying diode D₀ and a smoothing capacitor C₀. Connected between one extremity of the transformer secondary N₂ and the output terminal 11, the rectifying diode D₀ is so oriented as to be conductive when the primary switch Q₁ is off, and nonconductive when the primary switch Q₁ is on. The capacitor C₀ is connected in parallel with the transformer secondary N₂ via the diode D₀. A unidirectional output voltage is thus obtained between the pair of output terminals 11 and 12 for feeding a load 20 connected thereto.

The present invention particularly features a soft-switching circuit 7 comprising an ancillary switch Q₂ and two ancillary diodes D_(a) and D_(q2), in addition to the noted transformer tertiary N₃. This transformer tertiary is so polarized that the current flowing therethrough may cause a voltage to be developed across the transformer primary N₁ with consequent discharge of the soft-switching capacitor C_(q1). The transformer tertiary N₃ has leakage inductance L_(a). An additional inductor, not shown, may be connected in series with the transformer tertiary N₃ in cases where the leakage inductance N_(a) fails to provide a required amount of inductance.

The transformer tertiary N₃, ancillary reverse-blocking diode D_(a) and ancillary switch Q₂ are connected in series with one another. This serial connection is connected in parallel with the smoothing capacitor C₁ and also with the serial connection of the transformer primary N₁ and primary switch Q₁. Shown as a field-effect transistor, the ancillary switch Q₂ has an inbuilt parallel diode D_(q2) which is so oriented as to permit flow of reverse current. The ancillary switch Q₂ is understood to possess parasitic capacitance between its drain and source. The ancillary diode D_(a) is so oriented as to be forward-biased by the voltage induced across the transformer tertiary N₃ during the nonconducting periods of the primary switch Q₁.

As shown also in FIG. 1, a switch control circuit 8 has inputs connected to the pair of output terminals 11 and 12 by way of conductors 13 and 14, respectively, and an output connected to the control input of the primary switch Q₁ by way of a conductor 15, and another output connected to the control input of the ancillary switch Q₂ by way of a conductor 16. It is understood that the switch control circuit 8 is additionally connected to the sources of both switches Q₁ and Q₂, such connections being not specifically indicated in the drawings because of their impertinence to the instant invention. The switch control circuit 8 puts out a primary switch control signal V_(g1) for on-off control of the primary switch Q₁, and an ancillary switch control signal V_(g2) for on-off control of the ancillary switch Q₂.

FIG. 2 is a more detailed illustration of the switch control circuit 8. Included is a output voltage detector circuit 21 which is connected to the pair of input conductors 11 and 12, FIG. 1, by way of the conductors 13 and 14 for putting out a voltage proportional to the output voltage V_(o) of this power supply. The output of the output voltage detector circuit 21 is connected to one input of a differential amplifier 22, the other input of which is connected to a reference voltage source 23. The output of the differential amplifier 22 is connected both to one input of a first comparator 25 and, via a level-setting voltage source 26, to one input of a second comparator 27. The other inputs of the comparators 25 and 27 are both connected to a sawtooth generator circuit 24. The output of the first comparator 25 is connected to the control terminal of the primary switch Q₁, FIG. 1, by way of the output conductor 15. The output of the second comparator 27 is connected to the control input of the of the ancillary switch Q₂ by way of the output conductor 16.

As required or desired, the differential amplifier 22 could be coupled photoelectrically to both comparator 25 and level-setting circuit 26. The output voltage detector circuit 21 could also be coupled photoelectrically to the differential amplifier 22.

Inputting the output voltage of the output voltage detector circuit 21 and the reference voltage from its source 23, the differential amplifier 22 puts out the difference V_(r1) therebetween for delivery to the first comparator 25. This first comparator 25 compares the incoming difference voltage V_(r1) with the sawtooth voltage V_(t) from the sawtooth generator 24, as indicated at (A) in FIG. 3 and puts out a series of duration-modulated switch control pulses V_(g1) shown at (B) in the same figure. The duration-modulated switch control pulses V_(g1) are delivered over the first output conductor 15 to the gate of the primary switch Q₁, making its on-off control accordingly.

The level-setting power supply 26 has a voltage set at −V_(d), such that the voltage V_(r2) on its output side is less than the differential amplifier output voltage V_(r1) by V_(d). The second comparator 27 compares this voltage V_(r2) with the sawtooth voltage V_(t), as indicated also at (A) in FIG. 3, and puts out another series of duration-modulated switch control pulses V_(g2), shown at (C) in FIG. 3, which are greater in duration than the primary switch control pulses V_(g2). The second series of duration-modulated switch control pulses V_(g2) are delivered over the second output conductor 16 to the gate of the ancillary switch Q₂.

The primary switch control pulses V_(g1) and ancillary switch control pulses V_(g2) produced as above by the switch control circuit 8 are shown also at (A) and (B) in FIG. 4. It will be observed from this figure that each ancillary switch control pulse V_(g2) rises, as at t₁, thereby initiating conduction through the ancillary switch Q₂ when the primary switch Q₁ is off. Each primary switch control pulse V_(g1) rises, as at t₂, thereby initiating conduction through the primary switch Q₁ shortly after the conduction of the ancillary switch Q₂. The time interval from t₁ to t₂ is so determined as to minimize switching loss when the primary switch Q₁ is turned on.

Both primary switch Q₁ and ancillary switch Q₂ are shown to turn off at the same moment, as t₅ in both FIGS. 3 and 4. This showing is not mandatory, however. The ancillary switch Q₂ could be turned off at any moment from t₄, when the current I_(q2) through the ancillary switch Q₂ drops to zero as at (F) in FIG. 4, and t₅, when the primary switch Q₁ goes off. For turning off the ancillary switch Q₂ at t₄, for instance, as indicated by the broken line at (B) in FIG. 4, a monostable multivibrator may be connected to the output of the second comparator 27 of the switch control circuit 8, as indicated by the dashed outline labeled 28 in FIG. 2. This MMV 28 may be caused to produce pulses each lasting as from t₁ to t₄ in FIG. 4.

Operation

In use of the FIG. 1 power supply the pair of a.c. input terminals 1 and 2 are to be connected to an unshown source of a.c. power, and the pair of d.c. output terminals 11 and 12 to the desired load 20. The smoothing capacitor C₁ will be charged to the desired d.c. voltage V_(c1) as the switch Q₁ is turned on and off by the switch control circuit 8. The resulting steady-state operation of this representative switching power supply will be discussed hereinbelow with reference to FIGS. 4 and 5 which show the voltage and current waveforms appearing in various parts of the FIG. 1 circuitry.

FIG. 5 is explanatory of how the invention achieves improvements in power factor and input waveform. At (A) in this figure are shown the series of switch control pulses V_(g1) applied by the switch control circuit 8 to the primary switch Q₁. The primary switch Q₁ will be turned on and off during each cycle T of the switch control signal consisting of one pulse, as from t₂ to t₃, and one space between such pulses, as from t₃ to t₄. The repetition rate of these switch control pulses V_(g1) is now assumed to be 20 kHz. It is also understood that the 50 Hz sinusoidal a.c. voltage V_(ac), (E) in FIG. 5, is now applied between the pair of a.c. input terminals 1 and 2.

As the primary switch Q₁ is repeatedly turned on and off, the amplitudes or peak values of the output current I₄ of the rectifier circuit 4 and the current I_(q1) through the primary switch Q₁ will change, as at (B) and (C) in FIG. 5, in conformity with the amplitude of the a.c. input voltage V_(ac). Thus the a.c. input current I_(ac) shown at (D) in FIG. 5 will closely approximate a sinusoidal wave, with consequent improvement in power factor and waveform. It is to be noted that the primary inductor L₁ is connected to the tap 10 of the transformer primary N₁. As a result, there will be no flow of the primary inductor current I_(L1) unless the rectifier output voltage V₄ becomes higher than the tap voltage due to the voltage V_(c1) across the smoothing capacitor C₁. The rectifier output current I₄ and a.c. input current I_(ac) are both shown to flow from t₁ to t₆ and from t₈ to t₉ at (B) and (D) in FIG. 5.

The operation of the FIG. 1 power supply exclusive of the soft-switching circuit 7 will be explained in some more detail. A current will flow through the path comprising the rectifier circuit 4, primary inductor L₁, second part N_(1b) of the transformer primary N₁, and primary switch Q₁ during the conducting periods T_(on) of the primary switch Q₁, as from t₂ to t₃ in FIG. 5. A current will flow at the same time through the path comprising the smoothing capacitor C₁, transformer primary N₁, and primary switch Q₁. The voltage developing across the transformer secondary N₂ during these primary switch conducting periods T_(on) will be oriented to reverse-bias the diode D₀, holding the same nonconductive. Energy will therefore be stored on the transformer 5 during the primary switch conducting periods T_(on), as well as on the primary inductor L₁.

On the other hand, during the nonconducting periods T_(off) of the primary switch Q₁, as from t₃ to t₅ in FIG. 5, both primary inductor L₁ and transformer 5 will release the energy they have stored during the previous primary switch conducting period T_(on), causing the flow of the current I₄ through the path comprising the rectifier circuit 4, primary inductor L₁, first part N_(1a) of the transformer primary N₁, and smoothing capacitor C₁. This capacitor C₁ will therefore be charged.

The current I₄ charging the smoothing capacitor C₁ during each primary switch nonconducting period T_(off) will diminish with time and become zero as at t₄ in FIG. 5. During these nonconducting periods T_(off), due to energy release from the transformer 5, there will be induced across the transformer secondary N₂ a voltage oriented to cause conduction through the diode D₀ of the rectifying and smoothing circuit 6. The capacitor C₀ and load 18 will both be powered through the diode D₀.

One cycle of switching operation, lasting as from t₂ to t₅ in FIG. 5, has now come to an end. The same cycle will repeat itself after t₅ when the primary switch Q₁ is closed again. The power supply output voltage V₀ may exceed a predefined limit in the course of such repetition of switching cycles. Thereupon the switch control circuit 8 will respond by shortening the primary switch conducting periods T_(on) to an extent necessary to return the power supply output voltage V₀ to normal. The switch control circuit 8 will also respond to an excessive drop in the power supply output voltage V₀, by making the primary switch conducting periods T_(on) longer until the output voltage returns to normal.

Reference is now invited to FIG. 4 again for a study of how the soft-switching circuit 7 functions for the soft switching of the primary switch Q₁. As will be noted from (A) and (B) in FIG. 4, which show the primary switch control pulses V_(g1) and ancillary switch control pulses V_(g2), both primary switch Q₁ and ancillary switch Q₂ are off before t₁ in FIG. 4. Therefore, as in the t₃-t₅ period in FIG. 5, the diode current I_(do) will flow as at (H) in FIG. 4, and the voltage V_(q1) across the primary switch Q₁ and the voltage V_(q2) across the ancillary switch Q₂ will be both held high before t₁ as at (C) and (E) in FIG. 4.

The ancillary switch Q₂ is shown turned on at t₁ by the ancillary switch control signal V_(g2), (B) in FIG. 4. The current I_(q2) will then flow as at (F) in FIG. 4 through the path comprising the smoothing capacitor C₁, transformer tertiary N₃, ancillary diode D_(a), and ancillary switch Q₂. The flow of the current I_(q2) through the transformer tertiary N₃ will cause a voltage to be developed across the transformer secondary N₂, such that the output rectifier diode D₀ will be reverse-biased. The current I_(d0) through the diode D₀ will become zero, as at (H) in FIG. 4, upon nonconduction of the diode D₀. The clamping of the transformer primary N₁ via the transformer secondary N₂ will thus be eliminated.

Due to the current flow through the transformer tertiary N₃, moreover, the transformer primary N₁ will develop a voltage that is oriented opposite to the voltage across the smoothing capacitor C₁. Then the soft-switching capacitor C_(q1) will discharge through the path comprising the transformer primary N₁, tertiary N₃, ancillary diode D_(a), an ancillary switch Q₂. The voltage V_(q1) across the primary switch Q₁ will diminish until it becomes zero at t₂, as at (C) in FIG. 4. Since the transformer tertiary N₃ connected in series with the ancillary switch Q₂ possesses inductance L_(a), the current discharged by the soft-switching capacitor C_(q1) will flow by the resonance of the capacitance of the capacitor C_(q1) and the inductance L_(a) of the transformer tertiary N₃. The current I_(q2) through the ancillary switch Q₂ will start rising from t₁, as at (F) in FIG. 4. It is thus seen that the zero-current switching of the ancillary switch Q₂ is accomplished at t₁, with little or no power loss when this switch is turned on.

Upon completion of discharge by the soft-switching capacitor C_(q1) at t₂, the currents I_(q1) and I_(q2) shown at (D) and (F) in FIG. 4 will start flowing along the path comprising the transformer tertiary N₃, first ancillary diode D_(a), ancillary switch Q₂, primary switch protection diode D_(q1), and transformer primary N₁, due to the liberation of the energy that has been stored on the inductance L_(a) of the transformer tertiary N₃. The current I_(q2) through the ancillary switch Q₂ will start dwindling at t₂, and so will the current through the primary switch protection diode D_(q1). The current I_(q1) is shown at (D) in FIG. 4 as the sum of the current through the primary switch Q₁ and the current through its protection diode D_(q1). However, this current I_(q1) will be collectively referred to as the primary switch current for simplicity.

The voltage V_(c1) across the smoothing capacitor C₁ will be impressed to the transformer primary N₁ at t₂ when the primary switch Q₁ is turned on or when the primary switch protection diode D_(q1) conducts. The energy stored on the inductance L_(a) of the transformer tertiary N₃ will be released through the path comprising the first ancillary diode D_(a), ancillary switch Q₂, and smoothing capacitor C₁ as well. The current I_(q1) through the primary switch protection diode D_(q1), that is, through the primary switch Q₁ will become zero at t₃. Since the primary switch protection diode D_(q1) is conductive from t₂ to t₃ in FIG. 4, the voltage V_(q1) across the primary switch Q₁ will be approximately zero during this period. The primary switch Q₁ may therefore be turned on at zero voltage during the t₂-t₃ period. The primary switch control signal V_(g1) is shown to go high at t₂ at (A) in FIG. 4. However, in consideration of possible fluctuations in the moment the primary switch Q₁ is turned on in practice, this switch may preferably be turned on midway between t₂ and t₃.

It must nevertheless be pointed out that the primary switch Q₁ may be turned on before t₂ when the voltage V_(q1) across the same becomes zero, and not earlier than t₁ when it starts dwindling. Switching loss will then diminish to an extent to which the voltage V_(q1) has dropped at the moment the primary switch is turned on.

Some reduction of switching loss is also possible if the primary switch Q₁ is turned on shortly after t₃. With the primary switch Q₁ held open at t₃, the resonance capacitor C_(q1) will start to be charged at that moment. But if the primary switch Q₁ is turned on while the voltage across this capacitor C_(q1) is still less than the voltage V_(q1) across the primary switch during its nonconducting periods, then a corresponding reduction of the switching loss will be realized. Broadly speaking, therefore, the primary switch Q₁ may be turned on at any moment after t₁ when the ancillary switch Q₂ conducts, provided that the voltage V_(q1) across the primary switch is less than that during the nonconducting period before t₁.

Since FIG. 4 is explanatory of what is taking place from t₁ to t₆ in FIG. 5, the primary switch protection diode D_(q1) is understood to conduct at t₂ in FIG. 4. Then the current I_(L1) through the first inductance coil L₁ will start increasing as at (G) in FIG. 4.

At t₃, when the primary switch protection diode D_(q1) becomes incapable of being held conductive, the current I_(q1) through the primary switch Q₁ will become zero and thereafter start flowing positive as at (D) in FIG. 4. Thus, during the ensuing t₃-t₄ period, the primary switch current I_(q1) will flow along both the path comprising the rectifier circuit 4, primary inductor L₁, second part N_(1b) of the transformer primary N₁, and primary switch Q₁, and the path comprising the smoothing capacitor C₁, transformer primary N₁, and primary switch Q₁.

It is understood that energy release from the inductance L_(a) of the transformer tertiary N₃ comes to an end at t₄, rather than at t₃, in this particular embodiment. For this reason, during the t₃-t₄ period, the current I_(q2) will flow as at (F) in FIG. 4 along the path comprising the transformer tertiary N₃, first ancillary diode D_(a), ancillary switch Q₂, and smoothing capacitor C₁. The rectifying diode D₀ will be reverse biased by the voltage building up across the transformer secondary N₂ when the primary switch current I_(q1) is going positive as from t₃ to t₄, and from t₄ to t₅, in FIG. 4. The current I_(do) through this diode D₀ will therefore remain zero as at (H) in FIG. 4, and the energy will be stored on the transformer 5.

The current I_(q2) through the ancillary switch Q₂ will be zero as at (F) in FIG. 4, whereas the current I_(q1) through the primary switch Q₁ will flow as at (D) in FIG. 4, from t₄ to t₅. During this period, as from t₃ to t₄, there will be current flow both along the first path comprising the first a.c. input terminal 1, filter 3, first diode D₁, inductor L₁, second part N_(1b) of the transformer primary N₁, primary switch Q₁, fourth diode D₄, filter 3, and second a.c. input terminal 2, and along the second path comprising the smoothing capacitor C₁, transformer primary N₁, and primary switch Q₁. The current on the first path is for improvements in power factor and input waveform and equivalent to the first inductance coil current I_(L1) shown at (G) in FIG. 4. The current on the second path is for d.c. to d.c. conversion. The current I_(d0) through the diode D₀ will be zero from t₄ to t₅ as at (H) in FIG. 4, so that energy will be stored on the transformer 5 during this period.

During the t₄-t₅ period, owing to the voltage across the transformer primary N₁, there will be induced across the transformer tertiary N₃ a voltage that is opposite in orientation to the voltage V_(c1) across the smoothing capacitor C₁. The ancillary diode D_(a) will therefore be nonconductive. The voltage V_(q2) across the ancillary switch Q₂ will be zero as at (E) in FIG. 4, and so will be the current I_(q2) through the ancillary switch Q₂. The ancillary switch Q₂ may therefore be turned off at any moment during this t₄-t₅ period for both zero-voltage and zero-current switching.

The conduction control of the ancillary switch Q₂ is shown to end at t₅ when that of the primary switch Q₁ also ends, in this particular embodiment of the invention. The zero-voltage and zero-current switching of the ancillary switch Q₂ is thus accomplished for reduction of switching loss when the ancillary switch is turned off. In practice, of course, the ancillary switch Q₂ may be turned off at t₄, as indicated by the broken line at (B) in FIG. 4, or at any other moment from t₄ to t₅.

As plotted at (D) in FIG. 4, the current I_(q1) through the primary switch Q₁ will drop to zero at t₅ when this switch is turned off. The current will flow instead into the soft-switching capacitor C_(q1) thereby charging the same. The voltage V_(q1) across the primary switch Q₁ will rise with a gradient as at (C) in FIG. 4. The zero-voltage turnoff of the primary switch Q₁ has thus been achieved. The current charging the soft switching capacitor C_(q1) as above will flow both along the path comprising the rectifier circuit 4, inductor L₁, second part N_(1b) of the transformer primary N₁, and soft switching capacitor C_(q1), and along the path comprising the smoothing capacitor C₁, transformer primary N₁, and soft switching capacitor C_(q1).

The inductor L₁ and transformer 5 will both release their energy during the ensuing t₆-t₇ period. The diode D₀ of the rectifying and smoothing circuit 6 will then be forward biased by the voltage across the transformer secondary N₂, so that the diode current I_(do) will flow as at (H) in FIG. 4. The smoothing capacitor C₁ will be charged through the current path comprising the rectifier circuit 4, inductor L₁, first part N_(1a) of the transformer primary N₁, and smoothing capacitor C₁.

As indicated at (G) in FIG. 4, the current I_(L1) through the inductor L₁ will drop to zero at t₇. The diode D₀ will stay conductive thereafter thanks to energy liberation by the transformer 5. The t₇-t₈ period is a repetition of the pre-t₁ period in FIG. 4. One cycle of operation comes to an end at t₈. Another similar cycle will restart at t₈ when the ancillary switch Q₂ is turned on again.

The advantages gained by this particular embodiment of the invention may be recapitulated as follows:

1. The primary switch Q₁ is turned both on and off at zero voltage, assuring less switching loss and higher efficiency.

2. The ancillary switch Q₂ is turned on at zero current and off at zero voltage and zero current, resulting in less switching loss due to this ancillary switch.

3. The current I_(L1) through the inductor L₁ changes in peak value with the amplitude of the a.c. input voltage V_(ac), with consequent improvements in a.c. input power factor and waveform. Such improvements in power factor and waveform are accomplished with the aid of the primary switch Q₁ in the d.c.-to-d.c. converter circuit comprising the smoothing capacitor C₁, transformer 5, primary switch Q₁, and rectifying and smoothing circuit 6. The objectives of improved power factor, improved waveform, and output voltage control are realized with the simple circuitry.

4. The ancillary circuit 7 for the soft switching of the primary switch Q₁ makes use of the winding N₃ which is incorporated with the transformer 5 as its tertiary, thereby avoiding too much increase in the size and cost of the power supply.

5. By reason of the connection of the inductor L₁ to the tap 10 on the transformer primary N₁, the current I_(L1) does not flow through this first inductor even if the primary switch is closed, unless the potential on the first rectifier output conductor 43 grows higher than that at the tap 10. The first inductor current I_(L1) does not flow for this reason during the t₀-t₁, t₆-t₈, and t₉-t₁₀ periods in FIG. 5. Although this may seem disadvantageous from the standpoints of waveform and power factor improvements, it should also be taken into account that power loss does not occur at the first inductor L₁ as long as there is no current flow therethrough. Higher efficiency may therefore be attained without sacrifice in waveform and power factor through adjustment of the tap position on the transformer primary N₁.

Embodiment of FIG. 6

This alternative form of switching power supply features a modified transformer 5 _(a) having an untapped primary N₁. The primary inductor L₁ is connected to the junction between transformer primary N₁ and primary switch Q₁, instead of to the tap 10 as in FIG. 1. The primary inductor L₁ is therefore coupled to the smoothing capacitor C₁ via the whole of the transformer primary N₁ and directly to the primary switch Q₁. All the other details of construction are as set forth above with reference to FIGS. 1 and 2.

Upon conduction of the primary switch Q₁ the current I_(L1) will flow through the path comprising the first rectifier output conductor 43, primary inductor L₁, primary switch Q₁ and second rectifier output conductor 45 even when the a.c. input voltage V_(ac) is low in amplitude, as from t₀ to t₁, t₆ to t₈, and t₉ to t₁₀ in FIG. 5. For this reason the FIG. 6 power supply is preferable to its FIG. 1 counterpart from the viewpoints of improvements in input waveform and power factor. Offsetting these advantages is higher power loss as a result of the flow of the current I_(L1) through the primary inductor L₁ practically throughout the complete cycle of the a.c. input voltage V_(ac). A compromise may be made by employing the FIG. 1 device where efficiency matters, and that of FIG. 6 where better input waveform and higher power factor are more important. A reduction of switching loss represents a feature common to both embodiments as they both incorporate the soft-switching circuit 7.

Embodiment of FIG. 7

Another preferred form of switching power supply shown here differs from the FIG. 1 device only in the connections of the soft-switching circuit 7. Similar in construction to its FIG. 1 counterpart, the soft-switching circuit 7 includes the serial connection of the transformer tertiary N₃, ancillary diode D_(a), and ancillary switch Q₂. This serial circuit is directly connected in parallel with the primary switch Q₁, “directly” because the transformer primary N₁ is not inserted in the parallel connection as in FIG. 1. The serial circuit above is connected, instead, in parallel with the smoothing capacitor C₁ via the transformer primary N₁.

The operation of the FIG. 7 embodiment exclusive of the soft-switching circuit 7 is similar to that of the FIG. 1 embodiment exclusive of its soft-switching circuit 7. The operation of the FIG. 7 soft-switching circuit 7 is essentially similar to that of FIG. 1. The operation of the FIG. 7 device is therefore explainable in reference to the waveform diagram of FIG. 4.

The ancillary switch current I_(q2) will start flowing along the path comprising the smoothing capacitor C₁, transformer primary N₁, transformer tertiary N₃, ancillary diode D_(a), and ancillary switch Q₂ at t₁ in FIG. 4 when the ancillary switch Q₂ is turned on. The current flow through the transformer tertiary N₃ will cause the transformer secondary N₂ to develop a voltage that is oriented to reverse-bias the diode D₀, causing nonconduction therethrough as at (H) in FIG. 4. The transformer primary N₁ will be released from clamping by the voltage V₀ across the capacitor C₀ via the transformer secondary N₂. The soft switching capacitor C_(q1) will start discharging along the path comprising the transformer tertiary N₃, ancillary diode D_(a), and ancillary switch Q₁. Consequently, as indicated at (C) in FIG. 4, the voltage V_(q1) across the primary switch Q₁ will start diminishing at t₁ and drop to zero at t₂. The zero-voltage switching of the primary switch Q₁ will thus be accomplished by turning the same on at t₂ or any moment from t₂ to t₃. Since the transformer tertiary N₃ has inductance L_(a), the current I_(q2) flowing through the ancillary switch Q₂ during the t₁-t₂ period will rise with a gradient as at (F) in FIG. 4.

At t₂, when the soft switching capacitor C_(q1) completes its discharge and when the primary switch Q₁ is turned on as at (A) in FIG. 4, the transformer tertiary N₃ will start liberating the energy that has been stored during the t₁-t₂ period. The current will flow both along the path comprising the ancillary diode D_(a), ancillary switch Q₂ and primary switch protection diode D_(q1), and along the path comprising the ancillary diode D_(a), ancillary switch Q₂, smoothing capacitor C₁ and transformer primary N₁. The ancillary switch current I_(q2) will start diminishing at t₂ and drop to zero at t₄, as at (F) in FIG. 4. A zero-voltage turnoff of the ancillary switch Q₂ is therefore possible from t₄ to t₅.

The primary switch Q₁ is shown turned off at zero voltage at t₅. As has been set forth in connection with the FIG. 1 embodiment, the soft switching capacitor C_(q1) has then been charged.

The soft-switching circuit 7 and primary inductor L₁ of the FIG. 7 switching power supply function just like their counterparts of the FIG. 1 device. The FIG. 7 embodiment gains the same advantages as that of FIG. 1.

Embodiment of FIG. 8

The FIG. 8 embodiment is similar to that of FIG. 7 except that the primary inductor L₁ is connected to the junction between transformer primary N₁ and primary switch Q₁ as in FIG. 6. The transformer primary N₁ is therefore untapped. The operation of the FIG. 8 power supply exclusive of the soft-switching circuit 7 is analogous with that of the FIG. 6 counterpart. The soft-switching circuit 7 is of the same construction and connections as that of FIG. 7, so that the zero-voltage switching of the primary switch Q₁ is possible as in FIG. 7.

Embodiment of FIG. 9

The FIG. 9 embodiment represents an addition of an ancillary charging circuit 30 and reverse-blocking diode D₅ to the FIG. 1 embodiment, all the other details of construction being as previously set forth in conjunction with FIGS. 1 and 2. The ancillary charging circuit 30 comprises a transformer quaternary N₄, a capacitor C₂, two diodes D₆ and D₇, and an inductor L₂. Connected between an additional output conductor 44 of the rectifier circuit 4 and the smoothing capacitor C₁, the ancillary charging circuit 30 provides a voltage for addition to the rectifier output voltage V₄ between the additional rectifier output conductor 44 and the grounded rectifier output conductor 45. The reverse-blocking diode D₅ is connected in series with the primary inductor L₁.

The transformer quaternary N₄ of the ancillary charging circuit 30 has one of its opposite extremities connected to both transformer primary N₁ and smoothing capacitor C₁. The other extremity of the transformer quaternary N₄ is connected to the additional rectifier output conductor 44 via a serial connection of the ancillary charging capacitor C₂, ancillary charging inductor L₂, and second ancillary charging diode D₇. The first ancillary charging diode D₆ is connected in parallel with the transformer quaternary N₄ via the ancillary charging capacitor C₂. The ancillary charging capacitor C₂ is connected between ancillary charging inductor L₂ and transformer quaternary N₄. The first ancillary charging diode D₆ has its anode connected to the junction between ancillary charging capacitor C₂ and ancillary charging inductor L₂. The additional rectifier output conductor 44, to which is connected the ancillary charging circuit 30 as above, is connected to the junction 48 between the diodes D₁ and D₃ of the rectifier circuit 4.

The transformer of the FIG. 9 device is generally designated 5 _(b). The transformer 5 _(b) is similar in construction to the FIG. 1 transformer 5 except for the addition of the quaternary N₄, which of course is electro-magnetically coupled to the other windings of the transformer.

The operation of the FIG. 9 device except the ancillary charging circuit 30 is akin to that of FIG. 1. The smoothing capacitor C₁ will be charged to the desired d.c. voltage V_(c1) as the primary switch Q₁ is turned on and off by the switch control circuit 8. The ancillary charging capacitor C₂ will be charged to a voltage V_(c2) by the voltage across the transformer quaternary N₄. The resulting steady-state operation of this switching power supply, particularly of its ancillary charging circuit 30, will be discussed hereinbelow with reference to FIGS. 10 and 11 which show the voltage and current waveforms appearing in various parts of the FIG. 9 circuitry.

FIG. 10 is similar to FIG. 5, being illustrative of how the invention achieves improvements in power factor and input waveform by the FIG. 9 embodiment. At (A) in this waveform diagram are shown the series of switch control pulses V_(g1) applied by the switch control circuit 8 to the primary switch Q₁. The primary switch Q₁ will be turned on and off during each cycle T of the switch control signal consisting of one pulse, as from t₂ to t₃, and one space between such pulses, as from t₃ to t₄. The repetition rate of the switch control signal V_(g1) is assumed to be 20 kHz. At (E) in FIG. 10 is shown the 50 Hz sinusoidal a.c. voltage V_(ac) being applied between the pair of a.c. input terminals 1 and 2.

As the primary switch Q₁ is repeatedly turned on and off by the switch control pulses V_(g1), the amplitudes or peak values of the output current I₄ of the rectifier circuit 4 and the current I_(q1) through the primary switch Q₁ will change, as at (B) and (C) in FIG. 10, in conformity with the amplitude of the a.c. input voltage V_(ac). Thus the a.c. input current I_(ac) shown at (D) in FIG. 10 will closely approximate a sinusoidal wave, with consequent improvement in power factor and input waveform. The rectifier output current 14 is shown at (B) in FIG. 10 as the current flowing through the junction 48 between the diodes D₁ and D₃ and is the sum of the current I_(L1) through the primary inductor L₁ and the current I_(L2) through the ancillary charging inductor L₂.

It is to be noted that the primary inductor L₁ is connected to the tap 10 of the transformer primary N₁ via the reverse-blocking diode D₅ in this FIG. 9 embodiment. As a result, there will be no flow of the primary inductor current I_(L1) or of the a.c. input current I_(ac) when the rectifier output voltage V₄ is less than the tap voltage due to the voltage V_(c1) across the smoothing capacitor C₁, as from t₀ to t₁, from t₆ to t₈, and from t₉ to t₁₀. The rectifier output current I₄ and a.c. input current I_(ac) are both shown to flow from t₁ to t₆ and from t₈ to t₉ at (B) and (D) in FIG. 10.

The FIG. 9 power supply operates in three different modes depending upon the instantaneous value of the a.c. supply voltage V_(ac) shown at (E) in FIG. 10. Let V_(a) be a first voltage value that is equal to the voltage between the tap 10 on the transformer primary N₁ and the grounded conductor 45 during the conducting periods of the primary switch Q₁, and V_(b) be a second voltage value that is equal to the voltage V_(c1) across the smoothing capacitor C₁. Then, in the first 180 degrees of the a.c. supply voltage V_(ac), the power supply will operate in First Mode during the t₀-t₁ and t₆-t₇ periods when the a.c. supply voltage V_(ac) is between 0 and first value V_(a), in Second Mode during the t₁-t₃ and t₅-t₆ periods when the voltage V_(ac) is between first value V_(a) and second value V_(b), and in Third Mode during the t₃-t₅ period when the voltage V_(ac) is higher than the second value V_(b).

Incidentally, each negative half-cycle of the a.c. supply voltage V_(ac), as from t₇ to t₁₀ in FIG. 10, is inverted into the same shape as that of each positive half-cycle, as from t₀ to t₇, as the a.c. supply voltage is rectified by the rectifier circuit 4. The noted three modes of operation repeat themselves during the negative half-cycles. Also, in FIG. 10, the a.c. supply voltage V_(ac) is shown to cross the second value level V_(b) at the termination of one conducting period T_(on) of the primary switch Q₁. This showing is by way of example only; in practice, the crossing moment may come at other than the end of each conducting period.

In First Mode of operation, as from t₀ to t₁ and from t₆ to t₇ in FIG. 10, the primary switch current I_(q1) will flow as at (C) in FIG. 10 along the path comprising the smoothing capacitor C₁, transformer primary N₁, and primary switch Q₁ each time this switch conducts. No energy release from the transformer 5 will occur on its output side during these periods because then the diode D₀ is nonive. Energy will therefore be stored on the transformer 5 _(b). The potential at the tap 10 on the transformer primary N₁ is now higher than that of the first rectifier output conductor 43, so that there will be no flow of current I_(L1) through the primary inductor L₁. There will be no flow of current I_(L2) through the ancillary inductor L₂, either, because the voltage V_(C1) across the smoothing capacitor C₁ is now higher than the rectifier output voltage V₄.

The energy that has been stored as above on the transformer 5 _(b) will be released when the switch Q₁ subsequently goes off, with the consequent current flow along the path comprising the transformer secondary N₂, diode D₀, and capacitor C₀. The load 10 will therefore be powered even though the a.c. supply voltage V_(ac) is now lower than from t₁ to t₆.

In Second Mode, as from t₁ to t₃ and from t₅ to t₆ in FIG. 10, the potential at the transformer primary tap 10 will be less than that of the first rectifier output conductor 43. There will therefore be a flow of current I_(L1) through the primary inductor L₁. When the primary switch Q₁ goes on, the current I_(L1) will flow along the path comprising the first rectifier output conductor 43, primary inductor L₁, reverse-blocking diode D₅, transformer primary second part N_(1b), primary switch Q₁, and ground-potential conductor 45. Current will also flow along the path comprising the smoothing capacitor C₁, transformer primary N₁, and primary switch Q₁. Thus the current I_(q1) now flowing through the primary switch Q₁, shown at (C) in FIG. 10, will be the sum of the currents flowing through the two paths just mentioned. The diode D₇ of the ancillary charging circuit 30 will be off in Second Mode because the smoothing capacitor voltage V_(c1) will be higher than the rectifier output voltage V₄.

When the primary switch Q₁ is off, as from t₃ to t₄ in FIG. 10, in Second Mode, on the other hand, the current I_(L1) will flow to charge the smoothing capacitor C₁, with energy release from the primary inductor L₁. There will also be a current flow through the diode D₀ of the rectifying and smoothing circuit 6 as a result of energy release from the transformer 5 _(b) and primary inductor L₁. The primary inductor current I_(L1) will decrease in magnitude with the progress of the energy release from the transformer 5 _(b) and primary inductor L₁.

In Third Mode, as from t₃ to t₅ in FIG. 10, the a.c. input voltage V_(ac) and the rectifier output voltage V₄ will be higher than the smoothing capacitor voltage V_(c1), so that both first and second ancillary charging diodes D₆ and D₇ will be conductive. Both primary inductor current I_(L1) and ancillary inductor current I_(L2) will then flow, as will be hereinafter explained in more detail with reference to FIG. 11.

When the primary switch Q₁ is on, as from t₀ to t₁ in FIG. 11, in response to one of the primary switch control pulses shown at (A) in this figure, the primary inductor current I_(L1) will flow as at (F) in FIG. 11 along the same path as in Second Mode. There will also be a current flow in the circuit comprising the smoothing capacitor C₁, transformer primary N₁, and primary switch Q₁. The primary switch current I_(q1), (E) in FIG. 11, is the sum of the primary inductor current IL₁, (F) in FIG. 11, and the current discharged by the smoothing capacitor C₁. Also, during the conducting period Ton of the primary switch Q₁, there will be obtained across the transformer quaternary N₄ a voltage V_(n4), (H) in FIG. 11, depending upon the ratio of the turns of the transformer primary N₁ and ancillary N₄. This voltage V_(n4) is oriented to forwardly bias the first ancillary charging diode D₆, so that current will flow in the closed circuit comprising the transformer quaternary N₄, ancillary charging capacitor C₂, and first ancillary charging diode D₆. The ancillary charging capacitor C₂ will be charged with the polarity indicated in FIG. 9, with the consequent development of a voltage V_(C2) across the same.

As will be noted from (G) in FIG. 11, there has been a sustained flow of current I_(L2) through the ancillary inductor L₂ during the nonconducting period of the primary switch Q₁ preceding the t₀-t₁ conducting period T_(on) of FIG. 11. This ancillary inductor current I_(L2) will gradually decrease in magnitude during the t₀-t₁ period because then the anode potential of the first ancillary charging diode D₆ will be higher than that during each nonconducting period T_(off) of the primary switch Q₁. The ancillary inductor current I_(L2) will flow during this conducting period T_(on) along the path comprising the first a.c. input terminal 1, filter 3, first rectifier diode D₁, second ancillary charging diode D₇, ancillary charging inductor L₂, first ancillary charging diode D₆, smoothing capacitor C₁, fourth rectifier diode D₄, filter 3, and second a.c. input terminal 2. The ancillary inductor current I_(L2) will increase in magnitude with the a.c. input voltage V_(ac).

Then, during the ensuing nonconducting period T_(off) of the primary switch Q₁, as from t₁ to t₂ in FIG. 11, the smoothing capacitor C₁ will be charged by the primary inductor current I_(L1) as in Second Mode, and the current I_(do) will flow through the diode D₀ of the rectifying and smoothing circuit 6 as at (C) in FIG. 11. Further, as the transformer secondary N₂ will be clamped by the voltage V₀ across the capacitor C₀, a voltage V_(N4) will develop across the transformer quaternary N₄ as at (H) in FIG. 11. The transformer quaternary voltage V_(N4) during the nonconducting period T_(off) will be opposite in polarity to that during the conducting period T_(on), reverse-biasing the first ancillary charging diode D₆. Since this transformer quaternary voltage V_(N4) during the nonconducting period T_(off) will be opposite in polarity to the smoothing capacitor voltage V_(C1), the potential at the right-hand extremity, as viewed in FIG. 9, of the ancillary charging inductor L₂ will be less than that during the conducting period.

The ancillary charging inductor current I_(L2) will rise gradually in magnitude during the nonconducting period T_(off). The ancillary charging inductor current I_(L2) will flow along the path comprising the first a.c. input terminal 1, filter 3, first rectifier diode D₁, second ancillary charging diode D₇, ancillary charging inductor L₂, ancillary charging capacitor C₂, transformer quaternary N₄, smoothing capacitor C₁, fourth rectifier diode D₄, filter 3, and second a.c. input terminal 2, charging the smoothing capacitor C₁ in so doing. The smoothing capacitor C₁ is charged by both primary inductor current I_(L1) and ancillary inductor current I_(L2). It will therefore be appreciated that the smoothing capacitor C₁ is charged to the voltage V_(C1) that is higher than if, as has been the case heretofore, it is charged only by the primary inductor current I_(L1). The current I₄ through the rectifier circuit 4, shown at (B) in FIG. 11, is the sum of the primary and ancillary inductor currents I_(L1) and I_(L2), (F) and (G) in FIG. 11.

Incidentally, FIG. 11 is meant purely to illustrate how the current and voltage signals in question change with time. Their amplitudes are shown simplified or idealized.

The FIG. 9 embodiment possesses the advantages as that of FIG. 1 resulting from the primary inductor L₁ and soft-switching circuit 7. The ancillary charging circuit 30 offers the following additional advantages.

1. The smoothing capacitor C₁ is charged not only by the primary charging network comprising the rectifier circuit 4, primary inductor L₁, reverse-blocking diode D₅, and transformer primary first part N_(1a), but by the ancillary charging circuit 30 as well. If the smoothing capacitor is to be charged in the FIG. 1 circuitry to the same voltage as heretofore, the current I_(L1) flowing through the primary inductor L₁ can be of smaller magnitude than in the prior art charging circuit. Not only can the primary inductor L₁ be reduced in size, but also power loss is lessened here for higher overall efficiency of the power supply. The primary inductor current I_(L1) may be reduced, of course, only to such a level that the desired improvements in input current waveform and power factor does not become unattainable. The ancillary charging circuit 30 has its own power loss. Yet the current flowing through this circuit for charging the smoothing capacitor C₁ is only of such magnitude that the resulting power loss is negligible. Altogether, the FIG. 9 switching power supply possesses a decisive advantage over the prior art in terms of efficiency in operation and compactness in size.

2. If the current flowing through the primary inductor L₁ for charging the smoothing capacitor C₁ is of the same magnitude as heretofore, on the other hand, then the smoothing capacitor will be charged to the voltage V_(C1) that is higher than heretofore by the amount charged by the ancillary charging circuit 30. Such higher smoothing capacitor voltage V_(C1) will be effective to restrict the peaks of the current flowing into the smoothing capacitor C₁ at or adjacent the peaks of the rectifier output voltage V₄, resulting in the reduction of the higher harmonics of the a.c. input current I_(ac).

3. The second ancillary charging diode D₇ functions to block reverse current flow from ancillary charging circuit 30 toward primary inductor L₁. The primary inductor current I_(L1) is reduced in magnitude in this respect, too, for further curtailment of power loss.

Embodiment of FIG. 12

The FIG. 12 power supply is similar to that of FIG. 9 except for the connection of the primary inductor L₁ to the transformer primary N₁. The primary inductor L₁ is connected via the reverse-blocking diode D₅ to the junction between transformer primary N₁ and primary switch Q₁, instead of to the tap 10 as in FIG. 9. The transformer 5, shown in FIG. 12 is equivalent to the FIG. 9 transformer 5 _(b) minus the tap 10. The relationship between primary inductor L₁ and transformer primary N₁ in FIG. 12 is akin to that in FIG. 6, so that the FIG. 12 embodiment combines the advantages of the FIGS. 6 and 9 embodiments.

Embodiment of FIG. 13

This switching power supply employs a modified ancillary charging circuit 30 _(a) in place of its FIG. 9 counterpart 30 but is identical with the FIG. 9 embodiment in all the other details of construction. The alternate ancillary charging circuit 30 _(a) has the first ancillary charging diode D₆ connected between ancillary charging inductor L₂ and transformer quaternary N₃, and the ancillary charging capacitor C₂ connected between ancillary charging inductor L₂ and smoothing capacitor C₁. All the other details of construction are as previously set forth in connection with the FIG. 9 ancillary charging circuit 30.

The voltage that builds up across the transformer quaternary N₄ during the conducting periods of the primary switch Q₁ is oriented to reversely bias the first ancillary charging diode D₆. There will therefore be no current flow through this diode D₆ that would charge the ancillary charging capacitor C₂. A voltage capable of forwardly biasing the first ancillary charging diode D₆ will develop across the transformer quaternary N₄ during the nonconducting periods of the primary switch Q₁, so that a closed circuit of the transformer quaternary N₄, ancillary charging capacitor C₂, and first ancillary charging diode D₆ will be completed for the flow of current ancillary charging the ancillary charging capacitor C₂.

If the voltage drop across the second ancillary charging diode D₇ is disregarded, the voltage between the input-side terminal of the ancillary charging inductor L₂ and the ground-potential conductor 45 is equal to the rectifier output voltage V₄. The voltage between the output-side terminal of the ancillary charging inductor L₂ and the ground-potential conductor 45 is equal to the difference between the voltage V_(c1) across the smoothing capacitor C₁ and the voltage V_(c2) across the ancillary charging capacitor C₂. Thus the voltage V_(L2) across the ancillary charging inductor L₂ is defined as:

V _(L2) =V ₄−(V _(c1) −V _(c2))=V ₄ −V _(c1) +V _(c2).

It is thus seen that the ancillary charging inductor current I_(L2) flows only when the sum of V₄ and V_(c2) is greater than V_(c1). As in the FIG. 9 embodiment, the smoothing capacitor C₁ is charged both by the current I_(L1) through the primary inductor L₁ and by the current I_(L2) through the ancillary inductor L₂, to the voltage V_(c1) that is higher than that of the FIG. 1 embodiment.

Embodiment of FIG. 14

The transformer 5 _(b) of the FIG. 13 switching power supply is replaceable by the transformer 5 _(c) of the FIG. 12 device. FIG. 14 is an illustration of the resulting device, in which the primary inductor L₁ is connected via the reverse-blocking diode D₅ to the junction between transformer primary N₁ and primary switch Q₁ as in FIG. 12. This device gains the same advantages as do the FIGS. 12 and 13 embodiments.

Embodiment of FIG. 15

The ancillary charging circuit 30 and reverse-blocking diode D₅ of the FIG. 9 embodiment are applicable to the switching power supply of FIG. 7 as well. FIG. 15 shows the resulting device, which gains the advantages of both FIGS. 7 and 9 embodiments.

Embodiment of FIG. 16

The FIG. 16 embodiment is similar to that of FIG. 15 except that the primary inductor L₁ is connected via the reverse-blocking diode D₅ to the junction between transformer primary N₁ and primary switch Q₁ as in the FIG. 12 embodiment. The transformer 5 _(c) with the untapped primary N₁ is of the same construction as its FIG. 12 counterpart 5 _(c). The resulting power supply with the soft-switching circuit 7 is similar to the FIG. 8 device, and the ancillary charging circuit 30 identical with that of FIG. 9, so that the FIG. 16 embodiment combines the advantages of the FIGS. 8 and 9 embodiments.

Embodiment of FIG. 17

The ancillary charging circuit 30 of the FIG. 15 embodiment is replaceable by its FIG. 13 counterpart 30 _(a), as illustrated as a further preferred embodiment of the invention in FIG. 17. This FIG. 17 embodiment combines the advantages of the FIGS. 13 and 15 embodiments.

Embodiment of FIG. 18

In FIG. 18 the primary inductor L₁ is connected via the reverse-blocking diode D₅ to the junction between transformer primary N₁ and primary switch Q₁, as in FIG. 16. The transformer primary N₁ is therefore untapped. All the other details of construction are as set forth above with reference to FIG. 17. Having the same major circuitry, including the soft-switching circuit 7, as that of FIG. 8, and the same ancillary charging circuit 30 _(a) as that of FIG. 13, this FIG. 18 embodiment combines the advantages of the FIGS. 8 and 13 embodiments.

Embodiment of FIG. 19

The ancillary charging circuit 30 _(a) of FIG. 13 is modifiable as shown at 30 _(b) in FIG. 19, which shows a further preferred embodiment of the invention that is similar in all the other respects to that of FIG. 13. The modified ancillary charging circuit 30 _(b) incorporates only the transformer quaternary N₄ and ancillary charging diode D₇. Connected between the second rectifier output conductor 44 and smoothing capacitor C₁ via the ancillary charging diode D₇, the transformer quaternary N₄ is assumed to possess leakage inductance.

During the nonconducting periods of the switch Q₁, there will develop across the transformer quaternary N₄ a voltage V_(n4) that is oriented to forwardly bias the ancillary charging diode D₇. Current will flow through the ancillary charging diode D₇, charging the smoothing capacitor C₁, only when the sum of the rectifier output voltage V₄ and the transformer quaternary voltage V_(n4) grows higher than the voltage V_(c1) across the smoothing capacitor C₁.

This FIG. 19 embodiment offers the same advantages as does that of FIG. 13 except for the lack of smoothing effects by the inductor L₂ and capacitor C₂ of the FIG. 13 ancillary charging circuit 30 _(a). Counterbalancing this shortcoming are the simplicity in construction and compactness in size of the ancillary charging circuit 3 _(b).

The FIG. 19 ancillary charging circuit 30 _(b) lends itself to use with the transformer 5, of FIGS. 12 and 14. The cathode of the reverse-blocking diode D₅ may then be connected to the junction between transformer primary N₁ and primary switch Q₁ instead of to the tap 10 on the transformer primary. As an additional modification of the FIG. 19 embodiment, the soft-switching circuit 7 may be connected in parallel with the primary switch Q₁ as in the FIG. 7 embodiment. In short the ancillary charging circuit 30 _(b) of FIG. 19 may be substituted for the ancillary charging circuit 30 or 30 _(a) of FIGS. 9 and 12-18.

Embodiment of FIG. 20

The switching power supply is here shown comprising a modified rectifier circuit 4 _(a) and a modified ancillary charging circuit 30, in places of the rectifier circuit 4 and the ancillary charging circuit 30, respectively, of the FIG. 9 embodiment and is identical therewith in all the other respects. The modified rectifier circuit 4 _(a) has two diodes D₁₁, and D₁₂ in addition to the four noted diodes D₁-D₄. The fifth diode D₁₁ has its anode connected to the first a.c. input conductor 41, and its cathode to the additional rectifier output conductor 44. The sixth diode D₁₂ has its anode connected to the second a.c. input conductor 42, and its cathode to the additional rectifier output conductor 44. The second rectifier output conductor 44 is therefore supplied not with the outputs from the first and third diodes D₁ and D₃ but with the outputs from fifth and sixth diodes D₁₁, and D₁₂. These diodes D₁₁, and D₁₂ are substantially equal in electrical characteristics to the diodes D₁ and D₃, so that the voltage between second rectifier output conductor 44 and ground-potential conductor 45 is substantially equal to the voltage V₄ between first rectifier output conductor 43 and ground-potential conductor 45.

The modified ancillary charging circuit 30 _(c) is similar to the FIG. 9 ancillary charging circuit 30 except for the absence of the second ancillary charging diode D₇. The modified circuit 7 _(d) will nevertheless function just like the original circuit 7 as the two additional diodes D₁₁ and D₁₂ of the rectifier circuit 4 _(a) serve to block reverse current flow. It is understood that these additional diodes D₁₁ and D₁₂ are high-frequency diodes capable of responding to changes in the current through the ancillary inductor L₂ with the repeated conduction and nonconduction of the switch Q₁. The omission of the diode D₇ is inadvisable in cases where low-frequency diodes are used at D₁₁ and D₁₂.

The rectifier circuit 4 _(a) and ancillary charging circuit 30 _(c) of the FIG. 20 embodiment perform essentially the same functions as their FIG. 9 counterparts 4 and 30. The FIG. 20 embodiment thus wins the same advantages as that of FIG. 9. The rectifier circuit 4 _(a) could be used in the FIGS. 12-19 embodiments as well.

Possible Modifications

Notwithstanding the foregoing detailed disclosure, it is not desired that the present invention be limited by the exact showing of the drawings or the description thereof. The following, then, is a brief list of possible modifications or alterations of the illustrated embodiments which are all considered to fall within the scope of the invention:

1. All the ancillary charging circuits 30, 3 _(a), 30 _(b) could do without the second ancillary diode D₇. In the absence of this second ancillary diode D₇ the a.c. input current I_(ac) would flow during the t₀-t₁, t₆-t₈ and t₉-t₁₀ periods of FIG. 10 as well.

2. A high-frequency capacitor, with a capacitance less than that of the smoothing capacitor C₁, could be connected between the rectifier output conductors 43 and 44 in all the embodiments disclosed. In the FIG. 1 circuitry, for instance, the current flowing along the path comprising the primary inductor L₁, reverse-blocking diode D₅, transformer primary first part N_(1a), smoothing capacitor C₁, and rectifier circuit 4 during the non-conducting periods of the primary switch Q₁ could be caused to bypass the rectifier circuit if such a high-frequency capacitor were incorporated. The result would be the elimination of noise due to the diodes D₁-D₄.

3. An autotransformer could be used in places of the transformers 5, 5 _(a)-5 _(c).

4. All the switching power supplies could do without the transformer secondary N₂, and the rectifying and smoothing circuit 6 could be connected in parallel with the primary switch Q₁ to provide a step-up power supply.

5. The reverse-blocking diode D₅ could be connected between first rectifier output conductor 43 and primary inductor L₁, or omitted in cases where reverse current flow would present no problem.

6. An insulated-gate bipolar transistor or any other suitable semiconductor switches could be used in place of the FET switch Q₁.

7. The invention could be applied to “forward” switching power supplies in which the diode D₀ on the output side of the transformer conducts during the conducting periods of the primary switch Q₁.

8. The soft-switching circuit 7 could be connected to a tap 10 _(a) on the transformer primary N₁, as indicated by the broken lines in FIGS. 1, 6, 9, 12-14, 19 and 20, instead of to one extremity of the transformer primary. 

What is claimed is:
 1. A switching power supply capable of translating a.c. voltage into d.c. voltage, comprising: (a) a pair of a.c. input terminals for inputting a.c. voltage having a known frequency; (b) a pair of d.c. output terminals for outputting d.c. voltage; (c) a rectifier circuit connected to the pair of input terminals for rectifying the a.c. input voltage, the rectifier circuit having a first and a second output; (d) a transformer connected to the pair of a.c. input terminals via the rectifier circuit on one hand and, on the other hand, to the pair of d.c. output terminals, the transformer having a primary and an ancillary winding; (e) a rectifying and smoothing circuit connected between the transformer and the pair of d.c. output terminals for providing the d.c. output voltage; (f) a smoothing capacitor connected between a first extremity of the primary winding of the transformer and the second output of the rectifier circuit; (g) an inductor connected between the first output of the rectifier circuit and the smoothing capacitor via at least part of the primary winding of the transformer; (h) a primary switch connected between a second extremity of the primary winding of the transformer and the second output of the rectifier circuit; (i) soft-switching capacitance means associated with the primary switch; (j) an ancillary switch connected in parallel with the smoothing capacitor via at least the ancillary winding of the transformer for supplying to the ancillary winding a current of sufficient magnitude to cause the primary winding of the transformer to develop a voltage that enables the soft-switching capacitance means to discharge; and (k) a switch control circuit connected to the primary switch for on-off control of the primary switch at a repetition frequency higher than the frequency of the a.c. input voltage, the switch control circuit being also connected to the ancillary switch in order to initiate conduction through the ancillary switch earlier than the beginning of each conducting period of the primary switch and to terminate conduction through the ancillary switch not later than the end of each conducting period of the primary switch.
 2. The switching power supply of claim 1 wherein the ancillary switch is connected in parallel with the smoothing capacitor via the ancillary winding of the transformer and at least part of the primary winding of the transformer.
 3. The switching power supply of claim 1 further comprising a diode connected in series with the ancillary winding of the transformer and with the ancillary switch in order to prevent reverse current flow.
 4. The switching power supply of claim 1 wherein the rectifier circuit has a third output for putting out substantially the same rectifier output voltage between itself and the second output of the rectifier circuit as that between the first and the second output of the rectifier circuit, and wherein an ancillary charging circuit is provided which comprises a second ancillary winding of the transformer, the second ancillary winding being connected between the third output of the rectifier circuit and the smoothing capacitor and electromagnetically coupled to the primary winding of the transformer for providing a voltage for charging the smoothing capacitor.
 5. The switching power supply of claim 4 wherein the ancillary charging circuit further comprises: (a) a second inductor connected between the third output of the rectifier circuit and the second ancillary winding of the transformer; (b) an ancillary charging capacitor connected between the second inductor and the second ancillary winding of the transformer; and (c) an ancillary charging diode connected in parallel with the serial connection of the ancillary charging capacitor and the second ancillary winding of the transformer.
 6. The switching power supply of claim 5 wherein the ancillary charging circuit further comprises a reverse-blocking diode connected in series with the second inductor.
 7. The switching power supply of claim 4 wherein the ancillary charging circuit further comprises: (a) a second inductor connected between the third output of the rectifier circuit and the second ancillary winding of the transformer; (b) an ancillary charging diode connected between the second inductor and the second ancillary winding of the transformer; and (c) an ancillary charging capacitor connected in parallel with the serial connection of the ancillary charging diode and the second ancillary winding of the transformer.
 8. The switching power supply of claim 7 wherein the ancillary charging circuit further comprises a reverse-blocking diode connected in series with the second inductor.
 9. The switching power supply of claim 4 wherein the ancillary charging circuit further comprises a reverse-blocking diode connected between the third output of the rectifier circuit and the second ancillary winding of the transformer.
 10. The switching power supply of claim 4 further comprising a reverse-blocking diode connected in series with the inductor.
 11. The switching power supply of claim 4 wherein the rectifier circuit comprises: (a) a first diode having a first electrode connected to one of the pair of a.c. input terminals, and a second electrode connected to the first and the third output of the rectifier circuit; (b) a second diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said one a.c. input terminal; (c) a third diode having a first electrode connected to the other of the pair of a.c. input terminals, and a second electrode connected to the first and the third output of the rectifier circuit; and (d) a fourth diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said other a.c. input terminal.
 12. The switching power supply of claim 4 wherein the rectifier circuit comprises: (a) a first diode having a first electrode connected to one of the pair of a.c. input terminals, and a second electrode connected to the first output of the rectifier circuit; (b) a second diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said one a.c. input terminal; (c) a third diode having a first electrode connected to the other of the pair of a.c. input terminals, and a second electrode connected to the first output of the rectifier circuit; (d) a fourth diode having a first electrode connected to the second output of the rectifier circuit, and a second electrode connected to said other a.c. input terminal; (e) a fifth diode having a first electrode connected to said one a.c. input terminal, and a second electrode connected to the third output of the rectifier circuit; and (f) a sixth diode having a first electrode connected to said other a.c. input terminal, and a second electrode connected to the third output of the rectifier circuit.
 13. The switching power supply of claim 1 wherein the primary winding of the transformer has a tap, and wherein the inductor is connected between the first output of the rectifier circuit and the tap of the transformer primary winding.
 14. The switching power supply of claim 1 wherein the inductor is connected between the first output of the rectifier circuit and a junction between the primary winding of the transformer and the primary switch. 